1. Field of the Invention
The present invention relates to systems using electromagnetic transponders, that is, transceivers (most often, mobile) capable of being interrogated in a contactless and wireless manner by a unit (generally fixed), called a read and/or write terminal. The present invention more generally relates to a read or read/write terminal of transponders which have no independent power supply. Such transponders extract the power supply required by the electronic circuits included therein from the high frequency field radiated by an antenna of the read/write terminal. The present invention applies to such terminals, be they terminals which only read the transponder data (for example, an electronic label), or read/write terminals, which are likely to modify data of the transponder (for example, a contactless smart card).
2. Discussion of the Related Art
Systems using electromagnetic transponders are based on the use of oscillating circuits including a winding forming an antenna, on the transponder side and on the read/write terminal side. These circuits are intended to be coupled by a close magnetic field when the transponder enters the field of the read/write terminal.
FIG. 1 very schematically shows a conventional example of a data exchange system between a read/write terminal 1 and a transponder 10.
Generally, terminal 1 is essentially formed of a series oscillating circuit formed of an inductance L1 in series with a capacitor C1 and a resistor R1, between an output terminal 2 of an amplifier or antenna coupler 3 and a reference terminal 4 (generally the ground). Antenna coupler 3 receives a high-frequency transmission signal generated by a modulator (not shown) belonging to a control and data exploitation circuit 5 comprising, among others, a modulator-demodulator and a microprocessor for processing the control signals and the data. Circuit 5 also comprises a quartz oscillator providing a high-frequency reference signal. This reference signal is, in some cases, used as the reference signal REF of a phase demodulator 6 (Δφ) having the function of demodulating a possible data transmission originating from transponder 10. Signal REF may, instead of being extracted from circuit 5, that is, directly from the quartz oscillator, be sampled from output terminal 2 of antenna coupler 3 (dotted lines 9, FIG. 1).
Transponder 10 essentially includes a parallel oscillating circuit formed of an inductance L2, in parallel with a capacitor C2 between two input terminals 11, 12 of a control and processing circuit 13. Terminals 11 and 12 are, in practice, connected to the input of a rectifying means (not shown), the outputs of which form the D.C. power terminals of the circuits internal to the transponder. These circuits generally include, essentially, a microprocessor, a memory, a demodulator of the signals that may be received from terminal 1, and a modulator for transmitting information to the terminal.
In the absence of a data transmission from the terminal to the transponder, the high-frequency excitation signal is only used as a power source.
The possible transmission of information from the terminal to the transponder is performed, for example, by modulating the amplitude of the remote-supply carrier.
The transmission of information from transponder 10 to terminal 1 is generally performed by modifying the load of oscillating circuit L2, C2, so that the transponder takes more or less power from the high-frequency magnetic field. This variation is detected, on the side of terminal 1, insofar as the amplitude of the high-frequency excitation signal is maintained constant. Accordingly, a power variation of the transponder translates as an amplitude and phase variation of the current in antenna L1. This variation is then detected, for example, by means of phase demodulator 6 of terminal 1. For this purpose, for example, demodulator 6 receives a desired signal UTI originating from a current-to-voltage conversion by means of a resistor R3 between an input terminal of demodulator 6 and ground 4. Resistor R3 converts to voltage the current measured in oscillating circuit R1, L1, C1 by means, for example, of a current transformer 7 series-connected with the oscillating circuit. Current transformer 7 has been symbolized in FIG. 1 by two windings 7′, 7″. Primary winding 7′ is in series with the oscillating circuit. Secondary winding 7″ is connected by a first terminal to ground and by a second terminal to the input terminal of phase demodulator 6.
To transmit data from the transponder to the terminal, a modulation stage (not shown) of the transponder is controlled at a so-called sub-carrier frequency (for example, 847.5 kHz) much smaller (generally with a ratio of at least 10) than the frequency of the excitation signal of the terminal's oscillating circuit (for example, 13.56 MHz). The load variation on the transponder side is generally performed by means of an electronic switch for controlling a resistor or a capacitor modifying the load of oscillating circuit L2-C2. The electronic switch is controlled at the sub-carrier frequency to periodically submit the transponder's oscillating circuit to an additional damping with respect to the load formed by its exploitation circuits 13.
In the sub-carrier half-periods where the transponder's electronic switch is closed, demodulator 6 detects a slight phase shift (a few degrees, or even less than one degree) of the high-frequency carrier with respect to reference signal REF. Output 8 of demodulator 6 then gives back a signal which is an image of the control signal of the transponder's electronic switch, which can be decoded to restore the transmitted binary data.
A problem which arises with conventional read/write terminals using a phase demodulator is that the frequency response of the phase demodulator exhibits, when two oscillating circuits are tuned on the remote supply frequency (13.56 MHz), a zero (that is, the output voltage becomes zero) at a frequency of the signal to be demodulated corresponding to this remote supply frequency.
This phenomenon is illustrated in FIG. 2, which schematically shows the response of a phase demodulator 6. FIG. 2 shows the shape of voltage V8 at the output of demodulator 6 according to the frequency of the carrier on which the phase shift is detected. As illustrated in this drawing, the voltage becomes zero for a frequency f0 which corresponds, for a given coupling coefficient, to the resonance frequency of oscillating circuit L2-C2 of the transponder (f=½π√{square root over (LC)}).
To solve this problem, the oscillating circuits are generally detuned so that the two oscillating circuits of the terminal and of the transponder are not both tuned on the frequency of the remote supply carrier.
However, a disadvantage resulting therefrom is that this adversely affects the transponder remote supply, and thus the system range. Indeed, the power received by the transponder is maximum when the two oscillating circuits of the terminal and of the transponder are both tuned on the carrier frequency.
Another problem is that the manufacturing tolerances of the capacitors used for oscillating circuits, in particular for capacitor C2 of the transponder, generally are on the order of 10%. Accordingly, the significance of these manufacturing tolerances results in having to, for security, slightly shift from frequency f0 of the carrier to guarantee a phase demodulation by the terminal.
Thus, a significant disadvantage of conventional systems is that a compromise must be made between the remote supply and the phase demodulation capacity of the terminal.
Further, this compromise is difficult to find since the position of the “gap” in the phase demodulator response varies according to the mutual inductance between oscillating circuits. This mutual inductance depends on the distance which separates antennas L1 and L2 from the terminal and the transponder, and thus on the relative position of the transponder with respect to the terminal upon transmission.
This variation is illustrated in FIG. 3, which shows, for several intervals between a transponder and a terminal, examples of voltage-vs.-frequency characteristics, where the voltage represents the remote supply voltage of the transponder, for example, voltage V2 across capacitor C2, and where f corresponds to the excitation frequency of the terminal's series oscillating circuit.
The different curves illustrated in FIG. 3 are drawn for oscillating circuits tuned on frequency f0, that is, oscillating circuits of the terminal and of the transponder which are both sized to have a resonance frequency corresponding to the remote supply carrier. Curves g1, g2, g3, g4, g5, and g6 indicate decreasing distances between the transponder and the terminal. In other words, curve g1 which exhibits a small dome centered on frequency f0 substantially corresponds to the system limiting range. The more the distance reduces, the more the peak formed by the voltage-vs.-frequency characteristic increases, as illustrated by curves g2, g3, and g4. Curve g4 illustrates the optimal coupling position, that is, the distance at which the coupling is optimized by a maximum remote supply amplitude received by the transponder at frequency f0. Short of this distance, if the transponder is brought closer still to terminal 1, the voltage amplitude exhibits a decrease (curve g5) which becomes stronger (curve g6) as the interval between inductances L1 and L2 decreases, voltage V2 then exhibiting maximum values for the frequencies surrounding frequency f0. As a result, by shifting the resonance frequency on which the oscillating circuits are sized with respect to the carrier frequency, the vertical operation axis of the system is displaced in the characteristic of FIG. 3 and, accordingly, the recovered remote supply voltage is decreased, at least for curves g1 to g4.
The combined problems of the phase gap in phase demodulators and of the variation of the position of this phase gap with respect to the distance between inductances, associated with the component manufacturing tolerances, make conventional systems unreliable.
A first solution would be to replace the phase demodulator with an amplitude demodulator. Indeed, the charge variation caused by the transponder on the terminal's oscillating circuit also translates as a slight amplitude variation which can then be detected by a measurement, either of the current in the terminal's oscillating circuit, or of the voltage across capacitor C1.
However, this solution only transposes the problem since the spectral response of an amplitude demodulator also exhibits a demodulation gap, that is, a frequency for which the voltage obtained at the demodulator output is zero. The significant variation of the resonance frequencies of oscillating circuits due to the capacitor manufacturing tolerances, associated with the significant variation of the position of the demodulation gap according to the coupling between oscillating circuits results in that, in practice, the use of an amplitude demodulator risks posing substantially the same problems as the use of a phase demodulator.
Document WO-A-9618969 describes a contactless transceiver system in which the reader uses both a phase demodulator and an amplitude demodulator, and selects the output signal exhibiting the best level.
However, the solution described in this document is not fully satisfactory. Indeed, it does not enable, in practice, modifying the choice between the phase demodulation and the amplitude demodulation, upon reception, since the signals demodulated by the two demodulators are not necessarily synchronous.
Further, the disadvantages linked to the manufacturing tolerances of oscillating circuits and their operation drift remain.